The ability of communication, navigation, and a wide variety of frequency and timing measurement equipment to perform properly is determined in part by their phase noise performance. A common aspect of their design, manufacture, calibration, and ultimate use is the phase noise introduced by such equipment. Other performance parameters are certainly present and important but are often relatively more simple to predict, calibrate and control than phase noise performance. Various techniques have been used to measure the phase noise over portions of the range covered by the technique to which this invention is directed but with much reduced accuracy and/or added complexity. A survey of previously available techniques is described in section III of Extending the Range and Accuracy of Phase Noise Measurements by F. L. Walls et al, published in the Proceedings of the Symposium on Frequency Control, June 1988. Section IV A of that publication also describes the performance of the present system and section IV B outlines the measurement sequence used with the present invention. The article is incorporated herein by reference. Chapter 7 of Infrared and Millimeter Waves, Vol. II, pp. 239-289, by A. L. Lance et al also describes previously available techniques of phase noise measurement.
Specifically, and to briefly summarize these publications, the single oscillator techniques described yield a good resolution (noise floor) over a rather restricted range of Fourier frequencies. In order to measure the phase noise of a 10 GHz oscillator from 0.01 HZ to 1 GHz from the carrier would require from 5 to 10 different delay lines or reference cavities. Some of the cavities probably would have to be cryogenic to obtain the necessary resolution to measure the best sources that are available. In order to have good resolution using delay lines, the radio frequency (RF) signals would have to be encoded on optical carriers and transmitted on optical fibers. This technique is expensive and still falls short of the resolutions available with the two oscillator techniques described below.
The currently known two oscillator techniques have the best resolution over the entire measurement range. A typical block diagram of an arrangement to measure the phase noise of a precision source is shown in FIG. 1. FIG. 1 shows the use of a double balanced mixer (DBM) 11 receiving an input from a device under test, such as a precision signal source, at J1 and a reference signal input at J2. As will be clearly understood, these inputs are interchangeable since it is the phase difference between them that is the subject of measurement of the system. Both of the signals are provided as inputs to a double balanced mixer (DBM) 11, which is connected in a manner to cause it to function as a multiplier, and outputs the sum and difference frequencies of the inputs. As is well understood in the art, a double balanced mixer can be made to perform different functions, including those of a multiplier, as here, or an amplitude modulator, as will be discussed below, depending upon the ports to which inputs are applied.
Since the frequencies of the inputs to the DBM differ only by the amount of phase change being measured, the sum and difference frequencies will be widely separated and the sum frequency signal can be easily filtered out by low pass filter 15. The output of filter 15 is then provided to a bus which is connectable to one or more amplifiers which may be of either a dc coupled type 16 or ac coupled type 17, 18 configuration and thence to spectrum analyzers optimized for various frequency bands. For purposes of calibration of the spectrum analyzers, a noise reference 19 can be injected onto the bus at 19A, as illustrated, in a manner well-understood in the art. Specifically, for measurements where either the device under test or the reference source is voltage tunable, the signal on the bus can be amplified or attenuated as needed, filtered and fed back to the voltage tunable unit to maintain 90 degrees phase shift between the signals at the mixer as shown at 13 of FIG. 1. This nominally occurs at zero dc volts out of the mixer. In this case, phase shifter 12 can be set to any value including zero. For measurements where the phase of neither signal is adjustable, for example, measurements on an amplifier, where the reference signal drives J.sub.2 and J.sub.1 through the amplifier, adjustable phase shifter 12 is essential to achieve the 90 degrees phase shift.
Arrangements such as that of FIG. 1 exploit the fact that if the two oscillators (e.g. the reference or carrier frequency and the modulation reference frequency) are permitted to beat slowly against each other, the slope of the resulting waveform at zero output voltage from the mixer will have a value, in units of volts per radian, which is substantially a constant. For small phase deviations from zero output, the sine of the phase angle (i.e., the dc output voltage is approximately linear with phase deviation and will be a measure of the phase noise angle-to-voltage conversion constant in volts per radian. The difficulty with this approach in the past has been due to the fact that there has been no precision technique for isolating the changes in the slope which were due to frequency or amplitude dependent amplitude characteristics of the measurement system. Specifically, there had been no way to achieve a correction for the lumped frequency and amplitude characteristics of the measurement system at a particular carrier frequency and, therefore, no way to derive a high accuracy value for a constant relating the output voltage to the actual value of the phase noise as a function of Fourier frequency offset from the carrier signal.
The essence of the calibration of such a system is to accurately determine the conversion sensitivity of the mixer for transforming small changes in phase into changes in voltage as a function of the frequency of the phase change. The basic difficulty encountered is that virtually any frequency dependent variation in phase or voltage in any part of the system will result in a change in the voltage output, as is well understood in the art, while only those associated with a change due to the input phase noise of the input carrier signal from the device under test and the input reference source are of interest and must be isolated for high-accuracy measurements. This is complicated by the fact that the mixer output impedance has both resistive and reactive components whose value depends on the drive levels and the mixer termination. It is usually possible to obtain a very accurate measurement of this phase conversion for small changes occurring at a rate of less than 10 kHz. By carefully terminating the mixer in resistive loads it is possible to extend measurements up to 40 MHz in some cases with somewhat reduced accuracy and signal-to-noise ratio. The extrapolation to larger Fourier frequencies is extremely difficult because the output impedance of the mixer is active and frequency dependent. While high frequency errors in amplitude of detected phase noise are due to frequency dependent variations in amplifier gain, low frequency errors are due to the phase locked loop. These facts, coupled with the inherent small variations in amplifier gain with input impedance, frequency, and temperature makes it virtually impossible to obtain this conversion coefficient with an accuracy equal to or better than 10% (1 dB) over a wide range of Fourier frequencies.
As a solution to this problem, one might envision running one input signal at a high level and sweeping the other input signal while maintaining its level so low as not to saturate the IF amplifiers following the mixer. Unfortunately the mixer output impedance is a function of the drive levels and running one source at a low level changes its characteristics. This invalidates the calibration at the level of accuracy desired. Another approach is to insert a phase modulator directly in one of the lines. Here again, the approaches that have been used in the past added a significant amount of noise. Many of them are relatively narrow band devices such as cavities, which require many diverse devices to measure a wide range of carrier frequencies.
Still another approach is to insert a reference frequency into the output of the mixer. This approach has considerable merit but is even more difficult to implement since the reference signal must be flat from under 0.01 Hz to over 1 GHz (11 decades in frequency).